Signal select in underground line location

ABSTRACT

A transmitter and receiver for performing a signal select algorithm are provided. A transmitter for providing a signal on a line to be located includes at least one direct digital synthesizer, the direct digital synthesizer producing two component frequencies in response to an input square wave signal; and a feedback loop providing the input square wave.

RELATED APPLICATIONS

This application claims priority to U.S. Provisional Application No.61/640,441, filed on Apr. 30, 2012, which is herein incorporated byreference in its entirety.

BACKGROUND

1. Technical Field

The present disclosure relates to detection of underground lines and, inparticular, to a signal select in underground line location.

2. Description of Related Art

Underground pipe and cable locators (often termed line locators) haveexisted for many years and are described in many issued patents andother publications. Line locator systems typically include a mobilereceiver and a transmitter. The transmitter is coupled to a targetconductor, either by direct electrical connection or through induction,to provide a current signal on the target conductor. The receiverdetects and processes signals resulting from the electromagnetic fieldgenerated at the target conductor as a result of the current signal,which can be a continuous wave sinusoidal signal provided to the targetconductor by the transmitter.

The transmitter is often physically separate from the receiver, with atypical separation distance of several meters or in some cases up tomany kilometers. The transmitter couples the current signal, whosefrequency can be user chosen from a selectable set of frequencies, tothe target conductor. The frequency of the current signal applied to thetarget conductor can be referred to as the active locate frequency. Thetarget conductor then generates an electromagnetic field at the activelocate frequency in response to the current signal.

Different location methodologies and underground environments can callfor different active frequencies. The typical range of active locatefrequencies can be from several Hertz (for location of the targetconductor over separation distances between the transmitter and receiverof many kilometers) to 100 kHz or more. Significant radio frequencyinterference on the electromagnetic field detected by the receiver canbe present in the environment over this range. Therefore, receivers ofline location systems have often included highly tuned filters topreclude interference from outside sources from affecting themeasurement of signals at the desired active locate frequency from thetarget conductor. These filters can be tuned to receive signalsresulting from electromagnetic fields at each of the selectable activelocate frequencies and reject signals resulting from electromagneticfields at frequencies other than the active locate frequencies.

In line location systems, the signal strength parameter determined fromdetection of the electromagnetic field provides basis for derivedquantities of the current signal (i.e., the line current in the targetedconductor), position of the line locator receiver relative to the centerof the conductor, depth of the conductor from the line locator receiver,and can also be used as the input to a peak or null indicator (dependingon the orientation of the magnetic field to which that the detector issensitive). All line location systems measure signal strength on one ormore measurement channels.

Often in a crowded underground utility environment of metallic pipes andcables, coupling of signals at the active locating frequency from thetarget conductor to other adjacent underground conductors can occur.These conductors (lines) are not intended to be tracked by the linelocation system, but coupling of currents from the target conductor tothose neighboring conductors through various means (resistive,inductive, or capacitive), termed “bleedover,” can lead a line locatorastray such that the operator of the line location system ceasestracking the targeted conductor (e.g., pipe or cable of interest) andinstead begins following an adjacent line.

In conventional receivers, it is nearly impossible to determine whetherthe receiver is tracking the targeted conductor or whether the receiveris erroneously tracking a neighboring conductor. In complicatedunderground conductor topologies, the effect of interference fromelectromagnetic fields resulting from bleedover currents in neighboringconductors can result in significant asymmetrical electromagneticfields, which is termed field distortion. Further, conventional systemsthat attempt to distinguish between the targeted conductor andneighboring conductors typically rely on transmission of phaseinformation from the transmitter, which may be located at such adistance from the receiver of the line locator that receiving suchinformation is impractical.

Therefore, there is a need for line location systems capable ofaccurately determining the signal strength parameter from the targetedconductor exclusive of neighboring conductors that may provide signalsthat are a result of inductive or capacitive coupling, using a signalgeneration and processing method that utilizes only the targetedconductor (pipe or cable) as the transmission medium.

SUMMARY

In accordance with some embodiments, a transmitter and receiver forperforming signal select in underground line location is provided. Atransmitter for providing a signal on a line to be located includes atleast one direct digital synthesizer, the direct digital synthesizerproducing two component frequencies in response to an input square wavesignal; and a feedback loop providing the input square wave.

A method of receiving a frequency modulated signal from an undergroundline includes measuring phase of two frequency separated signals;calculating a gradient of the phase dispersal function; and determiningan offset based on the gradient. Another method of receiving a signalfrom an underground line includes processing incoming signals from oneor more antennas; demodulating the signal select waveform; establishinga phase reference for a transmitter phase; accessing a differencebetween the phase reference and a measured phase to provide a measure ofsignal select.

These and other embodiments will be described in further detail belowwith respect to the following figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates operation of a line locator system according to someembodiments of the present invention.

FIG. 2 illustrates at a high level a transmitter implementing someaspects of signal select according to some embodiments of the presentinvention.

FIG. 3 illustrates a feedback control system for the transmitterillustrated in FIG. 1.

FIG. 4 illustrates a current feedback and phase control loop for thetransmitter illustrated in FIG. 1.

FIG. 5 illustrates a transmitter with a signal select mode according tosome embodiments of the present invention.

FIG. 6 illustrates a receiver according to some embodiments of thepresent invention.

FIG. 7 illustrates narrow bandwidth signal processing according to someembodiments of the present invention.

FIG. 8 illustrates an FM demodulator that can extract the signal selectphase according to some embodiments of the present invention.

FIG. 9 illustrates recovery of the FSK phase reference according to someembodiments of the present invention.

The drawings may be better understood by reading the following detaileddescription.

DETAILED DESCRIPTION

In the following description, specific details are set forth describingsome embodiments of the present invention. It will be apparent, however,to one skilled in the art that some embodiments may be practiced withoutsome or all of these specific details. The specific embodimentsdisclosed herein are meant to be illustrative but not limiting. Oneskilled in the art may realize other elements that, although notspecifically described here, are within the scope and the spirit of thisdisclosure.

This description and the accompanying drawings that illustrate inventiveaspects and embodiments should not be taken as limiting—the claimsdefine the protected invention. Various changes may be made withoutdeparting from the spirit and scope of this description and the claims.In some instances, well-known structures and techniques have not beenshown or described in detail in order not to obscure the invention.

Additionally, the drawings are not to scale. Relative sizes ofcomponents are for illustrative purposes only and do not reflect theactual sizes that may occur in any actual embodiment of the invention.Like numbers in two or more figures represent the same or similarelements. Elements and their associated aspects that are described indetail with reference to one embodiment may, whenever practical, beincluded in other embodiments in which they are not specifically shownor described. For example, if an element is described in detail withreference to one embodiment and is not described with reference to asecond embodiment, the element may nevertheless be claimed as includedin the second embodiment.

Further, embodiments of the invention are illustrated with reference toelectrical schematics. One skilled in the art will recognize that theseelectrical schematics represent implementation by physical electricalcircuits, implementation by processors executing algorithms stored inmemory, or implementation by a combination of electrical circuits andprocessors executing algorithms.

FIG. 1 illustrates a line location system 100 according to someembodiments of the present invention. As shown in FIG. 1, line locationsystem 100 includes a transmitter 102 and a receiver 104. Transmitter102 is electrically coupled to a conductor 106 that is buried in theground 108. Conductor 106 may, for example, be a conducting pipe or awire and is generally considered to be a long conducting structure.Transmitter 102 provides an electrical signal along conductor 106, whichthen transmits an electromagnetic signal along its length. Theelectromagnetic signal is received by one or more antennae on receiver104. Receiver 104 is passed over the surface of the ground 108 in orderto locate the position of conductor 106 beneath the ground. From thesignal strength, the depth and position of conductor 106 can bedetermined.

Line location system 100, according to some embodiments of the presentinvention, includes a Signal Select system. Signal Select is a systemimplementation that exists to provide additional functionality to a linelocation system 100. Line location systems 100 can then employ theSignal Select system and use low frequency, alternating magnetic fieldsto perform a variety of remote sensing applications.

In some embodiments, the Signal Select system can use a frequency shiftkey (FSK) as a modulating function in transmitter 102, which allowsadditional information to be decoded by receiver 104. In particular,receiver 104 can decode the original phase of the transmitted signalregardless of any phase changes that are caused by the reactance(complex impedance) of the buried conductor 108.

The original Signal Select (disclosed in U.S. Pat. No. 6,411,073, anddeveloped further in U.S. Pat. No. 7,057,383, both of which are hereinincorporated by reference in their entirety) used frequency modulation,for example frequency shift key modulation, and provided a mechanism forthe receiver to decode the original transmitter phase. Subsequentdevelopments, by allowing for a real-time measurement of the ‘signaldistortion’ or ‘current bleed-off’ due to the reactance of conductor108, have been developed.

Some embodiments of line location system 100 according to the presenthave further improved on the Signal Select technology. In particular,some embodiments remove the production calibration process fortransmitter 102 and receiver 140, saving cost and time by utilizing asimpler process for manufacturing the system. Further, in someembodiments the architecture of transmitter 102 can be considerably lesscomplex than in previous systems, saving costs and improvingreliability. Further, in some embodiments receiver 104 need not rely onphase-locked loops (whether implemented by processors executing softwareor by electrical circuitry), providing for a faster response (lock-intime) for implementation of demodulation functions. In some embodiments,the overall signal-to-noise performance of line location system 100 canbe improved.

FIG. 2 illustrates generally the system architecture of transmitter 102.As shown in FIG. 2, a user interface 202 is coupled to a feedbackcontrol 204. The output of feedback control 204 is coupled through anamplifier 206 to a line 106 (pipe or cable). The line is terminated witha terminator 208. A current feedback 210 couples an input of thefeedback control 204 to the output signal from amplifier 206.

In some embodiments, the Signal Select system utilizes a compositewaveform, for example having 8 cycles at a lower frequency (f0*50/51)and 8 cycles at a higher frequency (f0*50/49). This bifurcatingwaveform, which is produced by transmitter 102, gives rise to amodulating function that operates on the carrier frequency (f0) with afrequency f0/16.

Some embodiments of transmitter 102 can utilize electronic hardware anda microprocessor to implement the Signal Select waveform as describedabove and be able to drive or induce a current into an attached pipe orcable, conductor 106. Conductor 106 can present almost any loadimpedance and therefore transmitter 102 accommodates phase shifts causedby the complex impedance (reactance) of conductor 106.

The embodiment of transmitter 102 illustrated in FIG. 2 illustrates somefeatures according to aspects of the present invention. Output amplifier206 illustrated in FIG. 2 can, for example, be a D-class amplifier thatis used in most standard cable locating systems. The example transmitter102 illustrated in FIG. 2 generally behaves as a constant currentgenerator such that the operator can request a fixed current at adefined carrier frequency through user interface 202.

In some embodiments of the invention, transmitter 102 utilizes DirectDigital Synthesizers (DDS) devices, which are used to form a closed loopfeedback system that may be under the control of a microprocessor. FIG.3 shows the functional components of some embodiments of feedbackcontrol 204. So as to avoid undue complication, the diagram illustratedin FIG. 3 avoids electronic detail, the subject of which is within theabilities of one skilled in the art. For example, some embodiments mayinclude three or more DDS devices and two or more microprocessors. Asillustrated in FIG. 3, the DDSs 304 and 312 can be utilized for drivingthe Signal Select waveform and correcting the phase of the Signal Selectmodulating function.

As shown in FIG. 3, DDS 304 can be a standard Direct Digital Synthesizersuch as an Analog Devices AD9833 DDS, for example. DDS 304 provides theSignal Select output signal 306 (denoted Vout in the drawing) that iscoupled to amplifier 206 illustrated in FIG. 2. The FSK signal at input308 is essentially a square wave at a frequency f0/16 and causes DDS 304to bifurcate between the two component frequencies (f0*50/51) and(f0*50/49)—the waveform having 8 cycles at each frequency. The FSKsignal is delayed by a delay 310 before being input to DDS 304.Accordingly, in response to the FSK signal, DDS 304 switches between thetwo values held in the frequency accumulator registers inside DDS 304.The other input to DDS 304 is the control phase signal at input 302,which is the feedback component. The control phase signal serves tophase advance or retard the Vout signal generated by DDS 304 and makesuse of a phase accumulate register inside DDS 304.

DDS 312 is a similar device as DDS 304 and uses the FSK control signalin an identical operation to that of DDS 304. DDS 312 does not, however,use the phase accumulate register as the purpose of DDS 312 is toprovide a fixed phase reference. Output 344, marked ‘sine-out’, is usedfor the purpose of counting the 8 cycles at the 2 component frequenciesand so for generating the FSK signal at input 308. As shown in FIG. 3,the signal from output 344 is input to a counter 314 that controls aswitch 316. Switch 316 switches between input 320 and input 318 andoutputs the FSK signal at input 308.

Outputs 342 ‘square (COS)’ and 340 ‘square (SIN)’ are in-phase andquadrature-phase square waves that allow feedback 204 to calculate thephase of the main current feedback signal that is connected to conductor106. As shown in FIG. 3, the signal from output 340 is mixed with thecurrent feedback signal from input 210 in multiplier 324. The outputsignal from multiplier 324 is input to filter 330. Similarly, the signalfrom output 342 is mixed with the current feedback signal from input 210in multiplier 326. The output from multiplier 326 is input to filter328. Filters 328 and 330 may, for example, be integrating filters andmay be first-order low-pass (LP) IIR filters. Such filters can beimplemented in hardware with op-amps or as a digital filter in thedomain of a microprocessor. The input signal at input 332, denoted ‘KInteg’, is essentially the time constant of the integrator (1/RC for ananalogue system). The output signals from filter 328 is combined ininverter 332 and summer 334 and the phase and amplitude is calculated inblock 336 to obtain an output signal at output 338. The output signal,marked ‘current phase’, can be utilized for the purpose of correctingthe problem of phase dispersal in some embodiments, as is discussedfurther below.

In some embodiments, feedback 204 can utilize topology around DDS 312that only uses a cosine feedback term from output 342. In thisimplementation the control phase at input 302 is adjusted until thecosine feedback signal is zero.

FIG. 4 shows a typical use of the current feedback 210 to control theoverall phase in DDS 304. In FIG. 4, the functions of DDS 304 and 312are combined into feedback block 402. As shown in FIG. 4, an integratorformed by adder 408 and delays 406 and 410 drives the ‘Control Phase’input. The input signal to this integrated is derives from the currentphase signal from output 338, which is summed with a dispersion offsetsignal from input 404 in summer 312 and amplified by amplifier 414. Thisintegrate signal from adder 408 phase advances or retards the entirewaveform. Other feedback processes can be utilized in some embodimentsof the invention.

Referring to the waveform definition of Signal Select as describedabove, the modulating function (a square wave with frequency f0/16)carries the original phase information of the transmitted waveformirrespective of any phase shifts which may arise as the signal travelsalong conductor 106.

FIG. 5 illustrates another example of transmitter 102 with a SignalSelect mode according to some embodiments of the present invention. Theexample of transmitter 102 shown in FIG. 5 can include multiplefunctional modes, as requested by the cable and pipe locating industry.As illustrated in FIG. 5, transmitter 102 can include a signalgeneration module 502, an output stage 526, and measurement circuitry518. A microprocessor 516 can be coupled to receive signals frommeasurement circuitry 518 and provide signals to signal generationmodule 502. Microprocessor 516 can be coupled to user interface 202 andcan provide multiple modes of operation according to the input receivedfrom user interface 202. Output stage 526 can receive an output signal(vout at output 306) and drives a load, for example conductor 106,according to the output signal. Signal generation module 502 and outputstage 526 provide signals for measurement circuitry 518.

As illustrated in FIG. 5, signal generation module 502 can include DDS504 and DDS 506. As is the case with DDS 304, DDS 504 provides theoutput signal that is coupled to conductor 106. DDS 506, similarly toDDS 312, provides a signal to measurement circuitry 518. Microprocessor516 provides control lines and an FSK control signal to DDS 504 and 506.A third DDS, DDS 508, is coupled to receive the control lines and FSKcontrol from microprocessor 516 and provide a reference signal tomicroprocessor 506 and measurement circuitry 518. As shown in FIG. 5,the output signal from DDS 506 can be a quadrature signal SIG_Q whilethe output signal from DDS 508 can be the in-phase signal SIG_I.

Output stage 526 can include a pulsed-wave modulation (PWM) generator510 coupled to receive the output signal from signal generation module502. The output signal from PWM generator 510 is input to an amplifier512, which may be a digital amplifier. The output signal from amplifier512 is input to output circuitry 514, which couples the signal to a loadsuch as conductor 106 and provides a feedback signal to measurementcircuitry 518.

Measurement circuitry 518 includes an amplifier 524 that receives thefeedback signal from output stage 526. The output signal from amplifier524 is input to a quadrature detector 522, which also receives theoutput signals SIG_Q and SIG_I from signal generation module 502.Quadrature detector 522 provides a detected quadrature signal DET_Q anddetected in-phase DET_I signal to a measurement processor 520. Theoutput signal from measurement processor 520 is provided tomicroprocessor 516.

When the Signal Select mode is chosen by the user from user interface202, microprocessor 516 will set DDS 504 in signal generation module 502to generate the proper frequency. At start-up the phase of this signalis set at a reference value.

In the same time DDS 506 and DDS 508 are set to generate the appropriatequadrature signals marked in FIG. 5 as “Sig_I” and “Sig_Q”. The SignalSelect generated by DDS 504 in signal generation module 502 is appliedto output stage 526 and from there into the load, which may be conductor106. Based on the load reactance, a phase offset between the voltage andthe current generated will be created. The measurement circuitry 518will sample the current in the load and apply it to a quadraturedetector 522, where also the above “Sig_I” and “Sig_Q” are injected. The“Det_I” and “Det_Q” signals at the output of quadrature detector 522 areapplied to a measurement processor 520 that will calculate the magnitudeand phase of the sampled signal. These results are sent tomicroprocessor 516 and based on that microprocessor 516 will control thelevel of the output signal and will adjust the phase of DDS 504accordingly, as requested by the Signal Select principle.

In the same time, because the Signal Select output signal is a FSK typeof waveform, the signal generated by DDS 508 in signal generation module502, respectively “Sig_I”, is used to trigger the switching of the twocomponents F1 and F2 of the FSK signal. This control signal is generatedby microprocessor 516 and applied to the signal generation module 502 asthe “FSK control” signal.

To improve the phase accuracy measurement specific for a highperformance Signal Select functionality, a dedicated software algorithmcan be executed by microprocessor 516, as specified above. Once thephase correction algorithm is executed the unit is ready for operation.In some embodiments, the signal generation module 502 is implementedusing a combination of three DDS circuits as illustrated in FIG. 5, butother circuitries that have frequency and phase controls can be used aswell.

It is desirable to be able to control the zero-crossing of the currentwaveform as it changes between the two component frequencies. Whentransmitter 102 is coupled to a complex impedance (typically havinginductance and capacitance), the modulating function utilizes a phaseoffset introduced to the demodulating device (in this case receiver 104)in order to determine the true original phase.

The two component frequencies discussed above will be shifted by anyreactance coupled to the output of transmitter 102 and importantly, thetwo component frequencies will be shifted by different amounts relativeto the carrier frequency ‘f0’. At the point of the transition, there isa discontinuity in the phase modulation, which can be corrected with acompensation dispersion offset as illustrated in FIG. 4. Someembodiments of the invention can use a principal of gradient measurementof the phase dispersal function (∂φ/∂f) to provide for the compensationdispersion offset in an adaptive manner. Accordingly transmitter 102takes measurements of the phase modulation φ firstly at a frequency‘f0−Δ/2’ and subsequently at a higher frequency ‘f0+Δ/2’. The frequencyspacing Δ is set at a level that results in high measurement accuracy asa function of phase contrast. A typical value for Δ can, for example, be30 Hz, although other values can be used. Once the phase dispersalfunction gradient (∂φ/∂f) is known, this value is added to the phasemodulating function in accordance with the following equations:

$\frac{\partial\varphi}{\partial f} = \frac{{\varphi \left( {f_{0} + \frac{\Delta}{2}} \right)} - {\varphi \left( {f_{0} - \frac{\Delta}{2}} \right)}}{\Delta}$${\varphi^{\prime} = {\varphi + {\alpha \; \frac{\partial\varphi}{\partial F}}}},$

where Φ′ is the compensation dispersion offset and α is a parameter thatrepresents the ‘run-time’ interpolation of the phase-dispersal gradient.In some embodiments, the parameter a can be chosen to be 25 Hz for thisprocess.

Receiver 104 performs a number of important functions. In particular,Receiver 104 may perform several functions related to the Signal Selectsystem described above with respect to transmitter 102. One suchfunction is that receiver 104 processes the incoming signals from a setof antennas, which may be ferrite antennas, and calculates an accuratesignal magnitude. A narrow bandwidth filter (for example about 7 Hz) canbe used in determining the signal magnitude. The incoming signalsdetected by the antenna are phase coherent but not phase locked toreceiver 102. The signal magnitude should be unaffected by thebifurcating frequency, which defines Signal Select signals according toembodiments of the present invention. Receiver 104 also demodulates theSignal Select waveform and establishes a phase reference for the truetransmitter phase—that is the phase before any subsequent line reactancecauses a phase shift. Further, receiver 104 also assesses the differencebetween the phase reference and the measured phase and in so doing canprovide a measure for the ‘signal distortion’.

FIG. 6 shows the principal elements of a receiver signal path 600 forany given antenna channel of receiver 104. In some embodiments, receiver104 can have multiple antennas and therefore multiple receiver signalpaths 600. As shown in FIG. 6, signal path 600 includes an antenna 602that receives an electromagnetic signal. The electromagnetic signal isfiltered in filter 604 and amplified in amplifier 606 prior to beingprocessed by a coder-decoder (CODEC) 608. CODEC 608 can be a Delta-Sigmadevice, which includes an intrinsic anti-alias filter. In the exampleshown in FIG. 6, CODEC 608 can sample, for example, 24-bit data attypically about 48 kHz. Therefore, the Signal Select signal can beprocessed entirely within the baseband of CODEC 608. The output signalfrom CODEC 608 is then input to a digital signal processor (DSP) 610.

FIG. 7 shows an example of DSP 610. DSP 610 can employ narrow bandsignal processing, which serves to compress the bandwidth from the CODECdata rate (48 kHz) to typically 94 Hz at the output of DSP 610.Correspondingly, the signal bandwidth is compressed from 24 kHz (outputfrom CODEC 608) to typically 7 Hz at the output from DSP 610. Thisbandwidth compression gives rise to an enhanced dynamic range and a goodreceiver would be expected to achieve 138 dBrms/√Hz from a 24-bit CODEC608 (assuming that CODEC 608 has a SINAD (signal to noise plusdistortion) ratio of >96 dB).

As shown in FIG. 7, DSP 610 receives the signal from CODEC 608 into twosignal processing paths, the paths processing in-phase and quadraturephase components, because there is no requirement for receiver 104 tolock to the phase of transmitter 102. This duplicated processing, asapplied to Signal Select, differs from previous implementations ofSignal Select because the previous implementations used a Phase LockedLoop to vector all the information into a single-phase carrier.

As shown in FIG. 7, in the in-phase path the signal from CODEC 608 isinput to multiplier 702, which mixes the output signal with an outputsignal from a numeric synthesizer 710. The output signal from multiplier702 is input to a synchronizer 704. The signal is then downsampled in aconverter 706, which can downsample for example by a factor of 512 asshown in FIG. 7, and filtered in a filter 708. Filter 708 may be alow-pass filter. Similarly, in the quadrature path the output signalfrom CODEC 608 is mixed with another output signal from numericsynthesizer 710 in multiplier 712. The output signal from multiplier 702is synchronized in synchronizer 714, down sampled in converter 716, andfiltered in filter 718, which can be a low pass filter. The in-phasepath and the quadrature path can include the same components, thedifference between the two paths being the signal from numericsynthesizer 710 that the output signal from CODEC 608 is mixed with.

Numeric synthesizer 710 can be similar to that described, for example,in U.S. Pat. No. 4,285,044 to Thomas et al (expired). Numericsynthesizer 710 provides sine and cosine outputs at the carrierfrequency ‘f₀’ and serves to shift the Signal Select waveform close to aDC signal. The exact programming carrier frequencies can be offset toaccount for the ‘average’ frequency which will be integrated in the FIRfilters 708 and 718 as follows:

${f_{0}^{*} = {f_{0}\left( \frac{\gamma^{2} - 1}{\gamma^{2\;}} \right)}},$

where γ is a parameter which can be, for example, 50.

As shown in FIG. 7, the output signals from filters 708 and 718 arecombined in combiner 720. Calculation 722 provides two outputs 724 and726. The bargraph signal from output 724, which represents the signalmagnitude, is given by

MAG=√{square root over (I ² +Q ²)},

where I is the in-phase signal magnitude and Q is the quadrature signalmagnitude. Similarly, the phase is given by

${Phase} = {{{Tan}^{- 1}\left( \frac{I}{Q} \right)}.}$

The phase serves as the phase reference of the carrier frequency ‘f₀’.The phase demodulation process follows a similar algorithm. In this casethe in-phase ‘I’ and quadrature phase ‘Q’ information is extracted at afraction, typically ⅛, the bandwidth of CODEC 608 (equivalentdownsampling factor of 8 comparing to 512 as shown in FIG. 7). Themodulation information is extracted using a digital FM demodulator suchas that shown in FIG. 8.

In an ideal situation, without any unwanted interference, the outputsignal from the demodulator (the input signals to combiner 720 in FIG.7) would be a 16 Hz square wave (actually a slightly asymmetric squarewave having 8 cycles at the lower frequency and 8 cycles at the higherfrequency). Therefore, combiner 720 can recover the true transmitterphase—the frequency shift key FSK.

The next process is the recovery of the true transmitter phase—thefrequency shift key FSK. As shown in FIG. 8, inputs 802 and 804 receivethe output signals from filters 708 and 718, respectively, whichcorrespond to the signals I and Q. As shown in FIG. 8, input signal Ifrom input 802 is provided to digital filter 810, delay 808, and isprovided to multiplier 806 to calculate I². Similarly, input signal Qfrom input 804 is provided to digital filter 812, delay 814, andmultiplier 816 to calculate Q². The output signal from multiplier 816and multiplier 806 are added in adder 818 to obtain I²+Q², which isprovided to delay 820. The output signal I from delay 808 is mixed withthe filtered signal Q, Q′, output from digital filter 812 in multiplier822. The output signal Q from delay 814 is mixed with the filteredsignal I, I′, from digital filter 810 in multiplier 824. The outputsignals from multipliers 822 and 824 are input to summer 826 to provideI′Q−Q′I. The output signals from delay 820 and summer 826 are providedto divider 828 to obtain the output value (I′Q−Q′I)/(I²+Q²), whichprovides an error indication. Delays 810, 814, and 820 temporarily alignthe signals with the output signals from filters 810 and 812.

As such, FIG. 8 illustrates a frequency demodulator. FIG. 8 takes thesignal from a compressed bandwidth, for example the CODEC bandwidth/10,and gives an output that follows the modulating function, for example asquare wave which is oscillating at f0/16.

FIG. 9 illustrates further processing 900. As shown in FIG. 9, the inputsignal at input 902 is divided by the input 16 in input 904, along witha phase increment from input 908, are input to a Thomas oscillator 910,which provides sine and cosine wave functions. The output signals fromthe FM demodulator, the signals from filters 708 and 718, are multipliedby the sine and cosine outputs from ‘Thomas’ Oscillator 910, which isprogrammed at ‘f0/16’. As shown in FIG. 9, the output signals fromoscillator 910 are mixed with the frequency demodulated signal generatedby FIG. 8, the output signal from output 830, that is received on input912. As shown in FIG. 9, the signal input on input 912 is mixed withsignals from filters 708 and 718 in multipliers 914 and 916 and theresults combined in combiner 920. Each of the signals is then processedthrough synchronizer 920, downsampler 922, and digital filter 924 andgain adjusted in amplifier 928 before being summed in summer 932.Calculation block 930 then provides the magnitude and phase of thesignal. Output 934 then provides the phase signal, which can be used asa reference phase.

Referring to FIG. 9 it is worthwhile to note the down-sampling ratio is64 so that the overall down-sampling equates to the narrow band signalprocessing algorithm. Therefore in this example: 512 (narrow band signalalgorithm)=8 (downsampling before FM demodulator)*64.

The final processing stage is the comparison of the signal phase, theoutput signal from output 726, and the demodulated phase reference, theoutput signal from output 830. In addition various numerical errors areeliminated. These errors are caused by mathematical truncation in theprogramming of the Thomas oscillator, if left unprocessed they wouldcause a slow drift in the reference phase and render the Signal Selectuseless. The phase data is cast to an integer (16-bit) which is aconvenient mechanism to describe the natural modulus function as itwraps at 2^(π) radians:

φ_(Signal  Select) = φ_(NB  Phase) − 16 × φ_(Demod  Phase) − φ_(Oscillator  Correction)$\varphi_{{Oscillator}\mspace{14mu} {Correction}} = {\varphi_{f\; 0\mspace{14mu} {Oscillator}} - {16 \times \varphi_{\frac{f\; 0}{16}{Oscillator}}}}$

In the above equation, φ_(NB Phase) corresponds to phase output signal726 shown in FIG. 7 and represents the average phase of the twofrequency components of the signal select signal. The parameterφ_(Demod Phase) corresponds to the output signal on output 934 of FIG.9.

The correction factor φ_(Oscillator Correction) is a small correctionthat exists because mathematical truncation using 32 bit calculationmeans that the numeric “Thomas” oscillators running at f0 and f0/16 willnot keep a fixed phase relationship over time. The drift may be small,but is enough to make the Signal Select signal useless after a shorttime (for example a minute or two). This phase drift is measured “on thefly” by receiver 104 and the results used. The difference shown above,then, represents the difference in phase of the frequency outputs ofnumeric synthesizer 710 (which produces a signal with frequency f0) andoscillator 910 (which produces a signal with frequency f0/16).

The output is ‘φ_(Signal Select)’, which is further utilized in SignalSelect processing. As the signal phase varies along conductor 106, thisfunction will track the phase change with respect to the referencecomponent ‘φ_(Demod Phase)’. The typical use of Signal Select is toprovide a real-time measure of ‘signal distortion’.

One further task is the task of phase tracking. The Transmitter and theReceiver operate as entirely separate systems and do not share a commonphase reference or system clock. To solve this problem, Receiver 104automatically tracks to the Transmitter output signal. This is a subtlefeature implemented in embodiments of the present invention, as Receiver104 has to track to an entirely ‘imaginary’ frequency—Signal Selectbeing defined as 8 cycles at (f0*50/51) and cycles at(f0*50/49) and thetracking therefore finds the true average frequency f0—even though itnever happens in time. The subtle point of this, is that withouttracking, f0 still contains the natural frequency errors betweenreceiver 104 and transmitter 102 (typically clock errors caused bycrystal oscillators and of the order of 30 ppm)—without correction thiserror will cause significant inaccuracy in the components of the aboveequations.

The process of phase tracking is implemented on the Narrow Band SignalProcessing algorithm as shown in FIG. 7. Essentially the phase output isdifferentiated and the resulting signal is integrated. This valuesubsequently serves as a negative feedback frequency error and is usedto adjust the frequency accumulator in the Thomas Oscillator (denotednumerical synthesizer in FIG. 7).

In the preceding specification, various embodiments have been describedwith reference to the accompanying drawings. It will, however, beevident that various modifications and changes may be made thereto, andadditional embodiments may be implemented, without departing from thebroader scope of the invention as set for in the claims that follow. Thespecification and drawings are accordingly to be regarded in anillustrative rather than restrictive sense.

What is claimed is:
 1. A transmitter for providing a signal on a line tobe located, comprising: at least one direct digital synthesizer, thedirect digital synthesizer producing an output signal with two componentfrequencies in response to an input square wave signal; and a feedbackloop providing a feedback signal coupled with the at least one directdigital synthesizer that adjusts the two component frequencies.
 2. Thetransmitter of claim 1, wherein the at least one direct digitalsynthesizer includes a first digital synthesizer that produces an outputsignal from the input square wave signal and a control phase signal. 3.The transmitter of claim 2, wherein the at least one direct digitalsynthesizer includes a second digital synthesizer that provides a sinewave output for generation of the input square wave and signals forgenerating a current phase signal from a feedback signal.
 4. Thetransmitter of claim 3, wherein the current phase signal is mixed with adispersion offset signal and integrated to form the control phasesignal.
 5. The transmitter of claim 1, further including an amplifierthat couples the output signal to a load conductor.
 6. A transmitter,comprising: a signal generation module that generates an output signalhaving two frequencies, the signal generation module including at leastone direct digital synthesizer; an output stage, the output stagecoupling the output signal to a load conductor; a measurement circuitrythat receives a signal from the output stage and phase signals from thesignal generation module and providing a feedback signal; and amicroprocessor coupled to the signal generation module and themeasurement circuitry, the microprocessor controlling the signalgeneration module in response to the feedback signal.
 7. The transmitterof claim 6, wherein the at least one digital signal processor of thesignal generation module includes a first direct digital synthesizerthat provides the output signal based on a control phase signal and anFSK control signal, a second direct digital synthesizer that provides aquadrature signal based on the FSK control signal and the control phasesignal, and a third direct digital synthesizer that provides an in-phasesignal based on the control phase signal and the FSK control signal. 8.The transmitter of claim 7, wherein the output stage includes apulse-wave modulation generator that receives the output signal from thesignal generation module, a digital filter that receives a modulatedoutput signal from the pulse-wave modulation generator, and an outputcircuit that couples a filtered output signal from the digital filter tothe load conductor, the output circuit providing a signal to themeasurement circuitry.
 9. The transmitter of claim 8, wherein themeasurement circuitry includes a measurement amplifier coupled toreceive the signal from the output circuit, a quadrature detectorcoupled to receive an amplifier signal from the measurement amplifier,and a measurement processor that provides the feedback signal to themicroprocessor based on the quadrature detector signal from thequadrature detector.
 10. A receiver, comprising an antenna provides anantenna signal based on a detected signal; an analog filter that filtersthe antenna signal to generate a filtered signal; an amplifier thatamplifiers the filtered signal to generate an amplified signal; a CODECthat receives the amplified signal and provides a decoded signal; and adigital signal processor that analyzes the decoded signal to receive asignal select signal that includes two frequencies, the digital signalprocessor including a demodulator to demodulate and downsample thedecoded signal to receive an in-phase component and a quadraturecomponent; and a calculation that provides the intensity and phase of asignal formed from the in-phase component and the quadrature component.11. The receiver of claim 10, wherein the digital signal processorprovides a phase error function based on the in-phase and quadraturecomponents.
 12. The receiver of claim 10, wherein the digital signalprocessor provides a phase correction based on the in-phase andquadrature components.
 13. The receiver of claim 10, wherein the digitalsignal processor provides phase tracking.
 14. A method of receiving afrequency modulated signal from an underground line, comprising:measuring a phase of two frequency separated signals; calculating agradient of a phase dispersal function; and determining an offset basedon the gradient.
 15. A method of receiving a signal from an undergroundline, comprising: processing incoming signals from one or more antennas;demodulating the signal select waveform; establishing a phase referencefor a transmitter phase; accessing a difference between the phasereference and a measured phase to provide a measure of signal select.